Advance Power Electronics Design | Economics Write


48 V-to-1 V Transformerless Stacked Active Bridge
Converters with Merged Regulation Stage

Jianglin Zhu, and Dragan Maksimovic
Colorado Power Electronics Center

Department of Electrical, Computer and Energy Engineering
University of Colorado, Boulder, Colorado 80309

Email: [email protected]

ABSTRACT

Transformerless stacked active bridge (TSAB) convert-
ers are hybrid converters derived from switched capacitor
converters by addition of small inductors. TSAB converters
achieve the highest efficiency around a nominal conversion
ratio because of “soft” charging/discharging of all flying
capacitors, low rms currents and zero-voltage-switching of
power switches. This paper introduces new configurations
of TSAB converters with inductive filter as opposed to ca-
pacitive filter at the output port, which opens opportunities
to merge a regulation stage at the output. The approach
is applied to 48 V to 1 V point-of-load (PoL) conversion
by merging a 6-to-1 Dickson TSAB with a multi-phase
buck converter at the regulation stage, greatly reduced the
need for a bulky intermediate bus capacitor. Experimental
results are provided for a 48 V-to-1 V, 100 A prototype
consisting of a 6-to-1 TSAB operating at 100-125 kHz
using 120 nH inductors, and an off-the-shelf four-phase
buck regulating stage operating at 500 kHz. The prototype
has 91.5% peak efficiency at 25 A and greater than 85%
efficiency up to 90 A.

I. INTRODUCTION

Motivated by the reduction in distribution losses, there is an
increased interest in higher voltage, e.g. Vbus = 48 V, dc dis-
tribution in data center applications, which in turn highlights
the need for high step-down point-of-load (POL) conversion,
e.g. 48-to-1 V, power xPU’s (GPUs, CPUs, and TPUs) on
server boards. In a conventional design, a two-stage ap-
proach is adopted, with an intermediate bus voltage VIB (e.g.
VIB = 12 V), where the output stage is typically implemented
using readily available multi-phase buck regulators. Various
approaches have been pursued to improve the performance
of the front-end Vbus-to-VIB stage, including switched-tank
converters [1], [2], LLC converters [3], resonant SC converter
[4], and TSAB converters [5]. To further improve the power
density and efficiency performance, direct Vbus = 48 V to PoL
converter topologies have also been investigated, such as the
Sigma converter [6], and the DIHC hybrid converter [7], [8].

 

Vbus

c1

c2

c1

c2

 

c2

c1

c1

c2

vIB

C3

C5

C1

L4C4

L2C2

 

Vout

 

c2s

c1s

iIB

vQ9

vQ1
 

Lb1

Lbn

 

Vbus

c1

c2

c1

c2

 

c2

c1

c1

c2

vIB

C3

C5

C1

L4C4

L2C2

 

Vout

 

c2s

c1s

iIB

vQ9

vQ1
 

Lb1

Lbn

 

Lf

Cf

(b)

(a)

Q1

Q2Q3

Q4

Q5

Q6

Q7

Q8

Q9

Q10

Q1

Q2Q3

Q4

Q5

Q6

Q7

Q8

Q9

Q10

Fig. 1: Inductive-output 6-to-1 TSAB merged with a multi-phase buck
regulation stage: (a) with a small LC filter in between, and (b) with no
filter in between.

In transformer-isolated converters, core losses and con-
duction losses in the transformer can be significant at high
switching frequency and high turns ratio. Advanced techniques
such as matrix transformer [9], and “integration” of secondary
winding and rectifier [3] have been proposed to address such
challenges. Since galvanic isolation is not required, trans-
formerless solutions, such as hybrid converters, are potentially
advantageous as they offer additional benefits such as reduced
switch voltage stresses, as well as soft switching.

In a two-stage design, the front-end Vbus-to-VIB converter
may be operated as an unregulated “DC transformer” (DCX)978-1-7281-1842-0/19/$31.00 c©2019 IEEE

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VIN

c1

c2

c1

c2

 

c2

c1

c1

c2

vx

L4C4

L2C2

voutLdc

 

VIN

 

vout

(a) (b)

c2s

c1s

C5

C3

C1

C5

C3

C1

C4

C2

Fig. 2: Two 6:1 Dickson TSAB converters: (a) capacitive-output TSAB where
C2, C4 and the output filter capacitor are considered tree branches, and (b)
inductive-output TSAB where the tree branches are C1, C3, C5. The tree
branches are highlighted in bold in the two configurations.

converter with efficiency optimized at a nominal step-down
conversion ratio. A step further has recently been proposed
in [10], [11], where the operation of a front-end switched-
capacitor stage is “merged” with a follow-up buck regulation
stage. Compared with a decoupled two-stage solution, the
front-end and the back-end converters are coupled in oper-
ation so that filter components such as the intermediate bus
capacitors can be reduced or eliminated. A similar approach
is proposed in this paper based on merging a transformerless
stacked active bridge (TSAB) converter and a multi-phase
buck regulator.

TSAB converters [12] are non-isolated hybrid dc-dc con-
verters with characteristics similar to the isolated Dual-Active-
Bridge (DAB) converter [13]. TSAB converters can be de-
rived from switched-capacitor (SC) converters by addition
of small ac inductors to eliminate “hard” capacitor charg-
ing/discharging losses [12]. The operation and control are
similar to the DAB converter [14]: output can be continuously
regulated by simple phase shift control, inductor peak current
are close to minimum and ZVS operation can be achieved
for most of the switches. Compared to other types of hybrid
converters, such as resonant switched-capacitor converters [4],
[15], or switched tank converters [2], [16], [17], which operate
near resonance (fs/fr ≈ 1), TSAB converters are designed to
operate above resonance (fs/fr > 1) with trapezoidal near-
minimum RMS inductor currents, and zero-voltage-switching
of power devices [12]. Above 98.5% efficiency and flat ef-
ficiency curves have been experimentally demonstrated on a
4-to-1 Dickson-based TSAB [5], and a 3-to-1 ladder-based
TSAB converter [18].

Two variants of a merged TSAB/regulation-stage converter
proposed in this paper are shown Fig. 1. The front-end
stage, which is a 6-to-1 Dickson TSAB with an inductive
output filter, is merged with a standard multi-phase buck POL
regulator. The variant in Fig. 1(a) retains a very small LC
filter between the two stages, while the intermediate-bus filter
is completely eliminated in the variant shown in Fig. 1 (b).

The paper is organized as follows: steady-state operation
and soft switching TSAB converters with inductive output port
are described in Section II. An analysis of the merged configu-
rations shown in Fig. 1 is presented in Section III. Design of a
48 V-to-PoL, 100 A prototype is described in Section IV, along
with key experimental results, including operating waveforms
and efficiency curves. The prototype consisting of a 6-to-1
inductive-output Dickson TSAB operating at 100-125 kHz
using 120 nH ac inductors, and an off-the-shelf four-phase
buck regulating stage operating at 500 kHz has 91.5% peak
efficiency at 25 A and greater than 85% efficiency up to 90 A.
Section V concludes the paper.

II. INDUCTIVE-OUTPUT TSAB CONVERTERS

TSAB converters introduced in [5], [12], [18] employ one
or more ac inductors to ensure soft charging and discharging
of flying capacitors. The output port has a dc filter capacitor.
The ac inductors and phase-shift operation yield trapezoidal
current waveforms with low RMS currents, as well as zero
voltage switching. As an example, the 6:1 Dickson TSAB
converter is shown in Fig. 2(a). In this converter, each inductor
shares the output current equally, and switches are operated
with reduced voltage stress (Vout and 2Vout, respectively) and
ZVS is achieved for most of the switches at sufficiently high
load current [5]. In general, capacitive-output TSAB converters
are obtained by inserting ac inductors into the link branches
of the corresponding two-phase switched-capacitor converter.
If the links remain the same in each switched state, hard
charging/discharging loops consisting of capacitors only are
eliminated.

As discussed in [12], the designation of links and tree
branches in an SC converter is not necessarily unique. In the
6:1 Dickson SC converter, if the output branch Vout is treated
as a tree branch, the tree capacitors are C2 and C4 and the
link capacitors are C1, C3, and C5. Inserting ac inductors in
series with the link capacitors yields the 6-to-1 Dickson TSAB
shown in Fig. 2(a). If, however, the output branch Vout is
treated as a link, the link capacitors are C2 and C4, while the
tree capacitors are C1, C3, and C5. Two ac inductors, L2 and
L4 are inserted in series with C2 and C4, respectively, while
a dc filter inductor Ldc is in the output branch, as shown in
Fig. 2(b). It should be noted that the dc filter inductor Ldc
carries the dc output current and ideally does not withstand
any volt-seconds , so that the inductance can be very small, and
inductor losses can be negligible compared to the ac inductors
which carry large ac currents.

A. Steady-state operation

In the inductive-output TSAB of Fig. 2(b), there are four
switched states, as illustrated in Fig. 4. States 1 and 2 are
the power-delivering states, where the inductors withstand
zero volt-seconds; states 1’ and 2’ are the polarity-reversal
states where the ac inductor currents are flipping polarities.
Assuming operation at nominal output voltage, which is the
same as in the original SC converter, the dc filter inductor does
not withstand any volt-seconds, so a very small inductance can

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c1

c2

c1s

c2s

iL2
iL4

IIB/3

iLdc
IIB

Ts

Tφ

t

j 1 2 11’2′ 2 1’2′

Fig. 3: Ideal operating waveforms in the inductive-output 6:1 Dickson TSAB
converter in Fig. 2(b). c1/c2, c1s/c2s are complementary 50% signals with
phase shift TÏ•. Ideal operation assumes nominal operation above resonance
fs � fr. Dead times are ignored, which is why there is no ripple in the
output filter inductor current iLdc.

(a) State 1′

VIN

C5 VOUT
Ldc

C1

L2C2

C3
L4C4

C3
L2

C2
C1

VOUT
Ldc

C5

L4

C4

VIN

C5 VOUT
Ldc

C1

L2

C2

C3

L4

C4

C3

L2 C2

C1

VOUT
Ldc

L4 C4

C5

(b) State 2′

(c) State 1 (d) State 2

vx
vx

vxvx

Fig. 4: Four switched-states in the inductive-output TSAB of Fig. 2(b). States
1 and 2 are power-delivering states, and states 1’ and 2’ are polarity-reversal
states.

be employed. Trapezoidal ac inductor currents can be realized
by introducing phase shifts between the control signals of the
two legs of the full-bridge rectifier (Q1 − Q4). Four control
signals c1,c2 and c1s, c2s are required, and the corresponding
timing diagram is shown in Fig. 3.

The balanced flying capacitor voltages are:

Vout ≈ VC1 ≈ Vbus − VC5 (1)

By symmetry,
VC3 ≈ Vbus/2 (2)

By volt-seconds balance applied to L2 and L4,

VC2 =
VC1 + VC3

2
(3)

 

VIN

c1

c2

c1

c2

 

c2

c1

c1

c2

C3

C5

C1

L4C4

L2C2

c2s

c1s Q1

Q2
D3

D4

Q5

Q6

D7

Q8

D9

Q10

vout

 

VIN

c1

c2

c1

c2

 

c2

c1

c1

c2

C3

C5

C1

L4C4

L2C2

c2s

c1s Q1

Q2
D3
D4

Q5

D6

Q7

D8

Q9

Q10

voutLdcLdc

(a) (b)

Fig. 5: Zero voltage switching transitions during (a) dead-time before c2 turns
on, and (b) dead-time before c1 turns on.

VC4 =
VC3 + VC5

2
(4)

Similar to the capacitive-output TSAB converter shown in
Fig. 2(a), the average output current is controlled by the phase
shift:

ILdc =
Vbus

8Lacfs
ϕ(1 − ϕ) (5)

where Ï• = 2TÏ•/Ts, and Lac = L2 = L4. Similar to a DAB
converter, operating away from the nominal voltage results
in increased volt-seconds on inductors and consequently in-
creased RMS currents.

B. Zero voltage switching transitions

For the full-bridge switches Q1 − Q4, inductor current of
Ldc forward biases the body diodes of either Q1,Q2 or Q3,Q4
during dead-times, so zero voltage switching can always be
achieved. Fig. 5 illustrates a ZVS transition for top switches
Q6 − Q9: the body diodes of these switches are forward-
biased by the inductor currents during the dead-time, assuming
sufficiently large inductive energy storage, i.e., sufficiently
large load current. Similar analysis shows that for the left of
the switches Q5,Q10, ZVS cannot be achieved. Importantly,
the ZVS switches Q6 − Q9 also block the highest voltage
(2Vout), while the hard switched devices only block Vout.

III. INDUCTIVE-OUTPUT TSAB CONVERTER MERGED
WITH A REGULATION STAGE

This section describes how an inductive-output TSAB con-
verter can be used as a front-end stage in a high step-down
application such as 48 V-to-PoL conversion. The follow-up
PoL converter can be a standard multi-phase buck converter,
as shown in Fig. 1(a). Furthermore, the small intermediate-bus
LC filter can be completely eliminated as shown in Fig. 2(b).
In this variant, the buck inductor serve the function of the
output inductor Ldc.

Eliminating the LC intermediate-bus filter imposes two
constraints. First, higher RMS currents are induced in some of
the TSAB capacitors and switches. The switching frequency of
the buck converter is typically much higher than the switching

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frequency of the TSAB converter. As a result, the buck induc-
tor current flows through C1 and C5 alternatively in states
1 and 2, which increases the RMS currents in capacitors C1
and C5 and the TSAB corresponding switches. The inductor
currents iL2 ad iL4 however, remain essentially unaffected.

Additionally, to ensure balanced voltages in the TSAB
converter, switching frequency ratio between the TSAB and
the buck stages should be an even integer value in the converter
of Fig. 1(b). If a multi-phase buck stage is employed, as
shown in Fig. 1(b), interleaving yields an equivalent switching
frequency proportional to the number of phases. The RMS
currents supplied by C1 and C5 can then be much smaller.

IV. DESIGN AND EXPERIMENTAL RESULTS FOR A
48 V-TO-1 V PROTOTYPE

This section presents the design and the experimental
results for a 48V-to-1 V prototype based on the merged
TSAB/regulation-stage converter shown in Fig. 1.

A. 48V-to-PoL prototype design

The design process for the 6-to-1 TSAB in Fig. 1 is
summarized in this section.

LTM4680

L2 L4

LTM4680

Fig. 6: 48 V-to-PoL/100 A prototype including 6:1 inductive-output TSAB
with planar inductors followed by a four-phase buck regulator. L2,L4 are
plannar inductors with 4 mm in height.

1) Selection of TSAB L and C values: The choice of the
inductive impedance ZL = ωsL and capacitive impedance
ZC = 1/(ωsC) affects the TSAB RMS currents. Let Rp be
the series resistance in the tanks consisting of ac inductors
L2, L4 and link capacitors C2, C4, respectively. Assuming
constant Rp, the quality factor Q =

√
(ZLZC)/Rp and the

ratio k = fs/fr =
√
ZL/ZC determine the waveshapes of the

ac inductor currents. As can be seen from simulations, a high
Q (Q > 10) and low k (k < 1.5) result in near-sinusoidal
current waveforms, whereas a low Q (Q < 3) and high k
(k > 1.5) result in near-trapezoidal waveforms. The k and
Q values also affect the required value of the phase shift for
a given output current. A larger phase shift corresponds to a
larger circulating current.

Switches

Q1 − Q4,Q5, Q10 EPC2023

Q6 − Q9 EPC2020

Passive components

C1 48 µF

C3,C5 80 µF

C2,C4 72 µF

L2,L4 120 nH

fsw,tsab 100-125 kHz

fsw,buck 500 kHz

k 1.7

Q ≈ 6

Inductor design

Number of turns 1

Core PC95 ELT11x4

AC resistance 4 mΩ

TABLE I: Parameters in the 48-to-1 V, 100 A prototype

In the prototype design, k = 1.7, Q = 10 and fs = 100 kHz
are selected, which ensures quasi-trapezoidal ac inductor cur-
rents with less than 2% required phase shifts at full load. The
corresponding parameter values are listed in Table I.

2) Design of ac inductors: Two low-profile 120 nH ac
inductors are implemented as planar low-profile inductors
(4 mm in height). The core material is PC95. Low inductances
allows for a single-turn design, which eliminates ac copper
losses due to proximity effects. Since the air gap is on the
opposite side of the PCB winding, the losses due to fringing
field are also relatively small. At the switching frequency, the
ac resistance obtained by finite-element simulation is found
to be very close to the dc resistance. Inductor core losses are
relatively low compared to the copper losses due to the low
peak-to-peak flux density.

3) Loss modeling: Major losses in the TSAB stage include
switch losses (conduction, switching losses), magnetic losses
(ac copper loss and core loss), and capacitor ESR losses.

In this design with four phase interleaved buck follow-up
stage, the RMS currents in the TSAB switches depend on
the nature of the intermediate-bus filter. If a large LC filter
with cutoff frequency lower than twice the TSAB switching
frequency is used, the RMS currents are minimized. On the
other hand, if a small (or no) filter is used, the RMS currents
are increased. In this case, the conduction loss model must
take into account the input current ripple in the follow-up buck
regulation stage.

Regarding switching losses, note that ZVS is achieved for
most of the TSAB switches assuming sufficient load current.
For the hard switching devices, and for the soft switching
devices at light load, switching losses are estimated by taking
into account voltage/current overlap losses and Coss losses.

Inductor copper losses are estimated based on ac resistance
found by finite-element analysis as shown in Table I, while
core losses are estimated using the iGSE method [19],.

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4) Multi-phase buck converter: The multi-phase buck con-
verter used in the prototype is based on two off-the-shelf
LTM4680 components in parallel [20], to obtain a four-phase
configuration with current mode control.

B. Experimental results

A 48 V-to-1 V converter prototype capable of 100 A output
current is shown in Fig. 6, and the component values are listed
in Table I.

Switch node voltages and ac inductor currents in the TSAB
stage in the configuration shown in Fig. 1 (a) are shown in
Fig. 7. Dips in vIB in Fig. 7 is attributed to dead-time between
signal c1/c2 and c1s/c2s. For example. during the dead-time
of Q1/Q2, both switch operate in reverse conduction mode,
and vIB collapses.

vIB(10V/div)

vQ9(12.5V/div)

iL2(5A/div)

iL4(4A/div)

Fig. 7: Waveforms in the TSAB stage with an LC filter: Vbus = 48 V,
VIB = 7.84 V, IIB = 15 A, fs = 100 kHz, time division: 4 µs/div. vQ9
and vIB are as shown in Fig. 1(a).

Efficiencies are measured separately for the TSAB and for
the four-phase buck stage from 10 A to 90 A load current
for Vbus = 48 V, Vout = 1 V as shown in Fig. 8. The
TSAB efficiency remains above 97.5% across the measured
load range with peak efficiency of 98.2% at 40 A. The four-
phase buck stage is operated with the four phases interleaved,
and the peak efficiency is around 94% at 20 A. The resultant
system peak efficiency is 91.5% at 25 A, and drops to
85% at 90 A. A small filter inductor Lf ≈ 60 nH and
filter capacitors Cf ≈ 22µF is used for each buck module.
The peak efficiency is comparable with other 48 V-to-1 V
hybrid converter solutions, such as DIHC [21], and heavy-
load efficiencies are improved. The heavy-load efficiency is
comparable to or slightly higher compared to transformer-
based two stage solutions, such as the current doubler design
reported in [22]. Additionally, compared with other direct
48 V-to-1 V solutions, regulation is much simpler because
readily available off-the-shelf buck modules can be employed.
Estimated loss breakdown in the TSAB stage for 1 V/40 A
output is shown in Fig. 9. The load current of the TSAB stage
is around 5 A, with longer than minimum dead times required
to achieve ZVS. In the model, the switch conduction losses
includes Rds,on losses as well as losses due to the PCB trace
resistances, and reverse-conduction losses are related to the
voltage drops across GaN FETs during dead-times.

As explained in Section III, removing the LC filter in
Fig. 1(b) is advantageous for size reduction. Fig. 10 compares
measured efficiency in the no-filter configuration of Fig. 1(b)

20 40 60 80 100
Output Current [A]

86

88

90

92

94

96

98

100

Buck

System

TSAB

91.5%

E
ff
ic

ie
n
cy

[
%

]

Fig. 8: Measured efficiencies of the 6:1 TSAB stage and the 8:1 four-phase
buck converter. The system efficiency is shown for the configuration in
Fig. 1(a) at 48 V input and 1 V output.

and small LC filter configuration of Fig. 1 (a). The small LC
filter yields improvement in efficiency in this case because the
RMS current in the switches are reduced compared to the case
with no filter. In terms of operation, LC filter is not necessary.
The operating waveforms for no filters are shown in Fig. 11.
Even with no input capacitors, the buck input voltage (vIB)
ripple is less than 1 V. This is because the flying capacitors,
especially C1 and C5 serve as an effective filter for the input
current ripple.

Conduction losses
27%

Switching losses
35%

Reverse-conduction losses
28%

Lac losses
8%

Ldc losses
1%

ESR losses
1%

Fig. 9: Modeled loss breakdown for the 6:1 TSAB prototype with Vbus =
48 V input, 1 V output, at the output current of 40 A.

V. CONCLUSIONS

TSAB converters [5], [12], [18] employ small ac induc-
tors while operating at relatively low switching frequency,
and offer high efficiency at nominal conversion ratio due to
“soft” charging/discharging of all capacitors and zero-voltage-
switching of power devices. TSAB converters can be cascaded
with a standard multi-phase buck regulation stage to construct
48 V-to point-of-load voltage around 1 V for server and

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Fig. 10: Measured efficiency for the configuration with a small LC filter
(Lf =60 nH, Cf = 8µF ) in Fig. 1 (a) and the no-filter configuration in
Fig. 1(b). Operating conditions: Vbus = 40 V, Vout = 1 V, fs,T SAB =
125 kHz, fs,buck = 500 kHz.

vQ9(12.5V/div)

Buck switch node
(5V/div)

vIB(1V/div)

iL4(2A/div)

Fig. 11: Experimental waveforms in the no-filter configuration in Fig. 1(b):
Vbus=30 V, Vout = 1 V, Iout = 30 A, fs,T SAB = 125 kHz, fs,buck =
500 kHz. Time division: 2 µs/div. vQ9 and vIB are as shown in Fig. 1(b).

other applications. To further reduce the losses associated with
ac inductors, this paper presents a new configuration of the
TSAB converter with an inductive output which eliminates
one of ac inductors. A follow-up multi-phase buck regulation
stage can be merged with the inductive-output TSAB using a
small LC filter. Completely removing the intermediate filter
is also feasible with a slight trade-off in conduction losses. An
experimental prototype is constructed using a 6-to-1 inductive-
output Dickson TSAB operating at 100-125 kHz followed
by an off-the-shelf four-phase buck regulator operating at
500 kHz. The prototype has 91.5% peak efficiency at 25 A
and greater than 85% efficiency up to 90 A.

ACKNOWLEDGMENT
The authors would like to acknowledge Lockheed Martin

for supporting research reported in this paper.

REFERENCES
[1] Y. Li, X. Lyu, D. Cao, S. Jiang, and C. Nan, “A 98.55% efficiency

switched-tank converter for data center application,” IEEE Transactions
on Industry Applications, 2018.

[2] S. Jiang, S. Saggini, C. Nan, X. Li, C. Chung, and M. Yazdani,
“Switched tank converters,” IEEE Transactions on Power Electronics,
2018.

[3] M. H. Ahmed, A. Nabih, F. C. Lee, and Q. Li, “Low loss integrated
inductor and transformer structure and application in regulated llc
converter for 48v bus converter,” IEEE Journal of Emerging and Selected
Topics in Power Electronics, pp. 1–1, 2019.

[4] Z. Ye, Y. Lei, and R. C. Pilawa-Podgurski, “A resonant switched ca-
pacitor based 4-to-1 bus converter achieving 2180 w/in 3 power density
and 98.9% peak efficiency,” in Applied Power Electronics Conference
and Exposition (APEC), 2018 IEEE. IEEE, 2018, pp. 121–126.

[5] J. Zhu and D. Maksimović, “A family of transformerless stacked
active bridge converters,” in Applied Power Electronics Conference and
Exposition (APEC). IEEE, 2019.

[6] M. Ahmed, C. Fei, F. C. Lee, and Q. Li, “High-efficiency high-power-
density 48/1v sigma converter voltage regulator module,” in 2017 IEEE
Applied Power Electronics Conference and Exposition (APEC), March
2017, pp. 2207–2212.

[7] G. Seo, R. Das, and H. Le, “Dual inductor hybrid converter for point-
of-load voltage regulator modules,” IEEE Transactions on Industry
Applications, pp. 1–1, 2019.

[8] ——, “A 95%-efficient 48V-to-1V/10A vrm hybrid converter using
interleaved dual inductors,” in 2018 IEEE Energy Conversion Congress
and Exposition (ECCE), 2018, pp. 3825–3830.

[9] M. H. Ahmed, C. Fei, F. C. Lee, and Q. Li, “48-v voltage regulator
module with pcb winding matrix transformer for future data centers,”
IEEE Transactions on Industrial Electronics, vol. 64, no. 12, pp. 9302–
9310, 2017.

[10] J. Baek, P. Wang, S. Jiang, and M. Chen, “Lego-pol: A 93.1% 54v-1.5v
300a merged-two-stage hybrid converter with a linear extendable group
operated point-of-load (lego-pol) architecture,” in 2019 20th Workshop
on Control and Modeling for Power Electronics (COMPEL), June 2019,
pp. 1–8.

[11] J. Baek, P. Wang, Y. Elasser, Y. Chen, S. Jiang, and M. Chen, “Lego-
pol: A 48V-1.5V 300A merged-two-stage hybrid converter for ultra-
high-current microprocessors,” in 2020 IEEE Applied Power Electronics
Conference and Exposition (APEC), 2020, pp. 490–497.

[12] J. Zhu, R. Schuess, and D. Maksimovic, “General properties and syn-
thesis of transformerless stacked active bridge converters,” in 2019 20th
Workshop on Control and Modeling for Power Electronics (COMPEL),
June 2019, pp. 1–6.

[13] R. W. De Doncker, D. M. Divan, and M. H. Kheraluwala, “A three-
phase soft-switched high-power-density dc/dc converter for high-power
applications,” IEEE transactions on industry applications, vol. 27, no. 1,
pp. 63–73, 1991.

[14] J. Zhu and D. Maksimović, “Dynamic modeling of a hybrid switched-
capacitor-based converter with phase-shift control,” in 2018 IEEE 19th
Workshop on Control and Modeling for Power Electronics (COMPEL),
June 2018, pp. 1–6.

[15] S. R. Pasternak, M. H. Kiani, J. S. Rentmeister, and J. T.
Stauth, “Modelling and Performance Limits of Switched-Capacitor
DC-DC Converters Capable of Resonant Operation with a Single
Inductor,” IEEE Journal of Emerging and Selected Topics in Power
Electronics, vol. 6777, no. c, pp. 1–1, 2017. [Online]. Available:
http://ieeexplore.ieee.org/document/7987696/

[16] S. Jiang, S. Saggini, C. Nan, and X. Li, “Switched Tank Converters,”
IEEE Transactions on Power Electronics, vol. 34, no. 6, pp. 5048–5062,
2019.

[17] Y. He, S. Jiang, and C. Nan, “Switched tank converter based partial
power architecture for voltage regulation applications,” in Applied Power
Electronics Conference and Exposition (APEC), 2018 IEEE. IEEE,
2018, pp. 91–97.

[18] J. Zhu, R. Scheuss, and D. Maksimovic, “Ladder transformerless stacked
active bridge converters,” in 2019 IEEE Energy Conversion Congress
and Exposition (ECCE), Sep. 2019, pp. 151–156.

[19] K. Venkatachalam, C. R. Sullivan, T. Abdallah, and H. Tacca, “Accurate
prediction of ferrite core loss with nonsimisoidal waveforms using only
steinmetz parameters,” Proceedings of the IEEE Workshop on Computers
in Power Electronics, COMPEL, vol. 2002-Janua, no. June, pp. 36–41,
2002.

[20] “Dual 30A or single 60A module regulator with digital power system
management,” Available at https://www.analog.com/media/en/technical-
documentation/data-sheets/LTM4680.pdf (2020/09/30).

[21] R. Das, G. Seo, and H. Le, “Analysis of dual-inductor hybrid converters
for extreme conversion ratios,” IEEE Journal of Emerging and Selected
Topics in Power Electronics, pp. 1–1, 2020.

[22] “Lmg5200 gan 48v to 1v point of load evaluation module,” Available
at https://www.ti.com/tool/LMG5200POLEVM-10 (2020/09/30).

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